Wireless devices have been in use for many years for enabling mobile communication of voice and data. Such devices can include mobile phones and wireless enabled personal digital assistants (PDA's) for example. FIG. 1 is a generic block diagram of the core components of such wireless devices. The wireless core 10 includes a base band processor 12 for controlling application specific functions of the wireless device and for providing and receiving voice or data signals to a radio frequency (RF) transceiver chip 14. The RF transceiver chip 14 is responsible for frequency up-conversion of transmission signals, and frequency down-conversion of received signals. RF transceiver chip 14 includes a receiver core 16 connected to an antenna 18 for receiving transmitted signals from a base station or another mobile device, and a transmitter core 20 for transmitting signals through the antenna 18 via a gain circuit 22. Those of skill in the art should understand that FIG. 1 is a simplified block diagram, and can include other functional blocks that may be necessary to enable proper operation or functionality.
Generally, the transmitter core 20 is responsible for up-converting electromagnetic signals from baseband to higher frequencies for transmission, while receiver core 16 is responsible for down-converting those high frequencies back to their original frequency band when they reach the receiver, processes known as up-conversion and down-conversion respectively. The original (or baseband) signal, may represent, for example, data, voice, or video. These baseband signals may be produced by transducers such as microphones or video cameras, be computer generated, or transferred from an electronic storage device.
FIG. 2 illustrates an example transmit path through the transmitter core 20 to the antenna 18. As shown in FIG. 2, the transmit path may include a mixer 202 arranged to receive baseband signals from the baseband processor 12. The mixer is responsible for up-converting the baseband signals to a higher frequency using a local oscillator signal generated by a local oscillator 204. The transmit path may further include a filter 206 for removing baseband components and suppressing harmonics and a power amplifier 208 for amplifying power of the modulated signal. The components in the transmit path are not comprehensive and any person of skill in the art will understand that the specific configuration will depend on the communication standard being adhered to and the chosen architecture implementation.
A known passive CMOS (complementary-symmetry metal-oxide-semiconductor) mixer circuit 300 will now be described with reference to FIG. 3.
The baseband signals are analog signals generated by modulating a baseband carrier with data, in accordance with any known protocol.
The CMOS passive mixer circuit 300 receives differential baseband signals (VBBP, VBBM) from a baseband processor. The term ‘differential’ is used here to describe that the baseband signals (VBBP, VBBM) have the same amplitude and are substantially in opposite phase to each other, i.e., 180 degrees out of phase. The mixer circuit 300 includes n-type metal oxide semiconductor field effect (NMOS) transistors 302, 304, 306, and 308 that are arranged to receive the baseband signals VBBP and VBBM and differential local oscillator signals (VLOP, VLOM). The NMOS transistors 302, 304, 306, and 308 provide differential outputs VOP and VOM.
Whilst the CMOS passive mixer circuit 300 has been described with respect to NMOS transistors, those skilled in the art will understand that transistors 302, 304, 306, and 308 may be selected to be p-type metal oxide semiconductor field effect (PMOS) transistors.
In operation, the mixer circuit 300 up-converts the baseband signals (VBBP, VBBM) to a desired RF transmit frequency using the local oscillator signals (VLOP, VLOM).
For the passive mixer 300 to operate, the baseband signals are required to drive the passive mixer that has a load at the output with minimum distortion. Any distortion from the baseband processor will degrade the linearity of the passive mixer.
One of the known protocols for RF signaling uses complex in-phase (I) and quadrature phase (Q) signals, where each can be in differential formats.
International Publication WO 2010/025556 discloses an IQ passive mixer 400 with driver circuitry 430 which will now be described with reference to FIG. 4.
The differential baseband input signals for the I and Q paths are labeled VBBQP, VBBQM, VBBIP, and VBBIM. These baseband input signals are input into the driver circuitry 430.
The driver circuitry 430 comprises source follower NMOS transistors 440, 444, 448 and 452 connected to bias NMOS transistors 442, 446, 450 and 454. The gate terminals of source follower NMOS transistors 440, 444, 448 and 452 receive the baseband input signals VBBQP, VBBQM, VBBIP, and VBBIM. The gate terminals of bias NMOS transistors 442, 446, 450 and 454 receive a bias voltage VBIAS. The outputs of the source follower NMOS transistors 440, 444, 448 and 452 are passed through resistors 460, 462, 464, 466 before being provided to the IQ passive mixer 400.
The passive IQ mixer 400 comprises NMOS transistors 402, 404, 406, 408, 410, 412, 414, and 416 for the I/Q paths and supplied by local oscillator signals labelled VLOIP, VLOIM, VLOQP, and VLOQM.
The outputs of the passive IQ mixer 400, namely VOP and VOM, are up-converted frequency signals that can be used to drive an amplifier, for example power amplifier 208 through accoupling capacitors (not shown in FIG. 4), prior to transmission.
The LO signals (VLOIP, VLOIM, VLOQP, and VLOQM) are each a square waveform (SO2 duty cycle) from 0V to 1.2V and are designed to have low rise and fall times. This arrangement enables the omission of surface acoustic wave (SAW) filters that are traditionally used at the transmitter's output. Accordingly, this helps to minimize the number of required external components, the required board area, and hence reduces the overall cost of the chip.
Placing a capacitive load on the outputs of the IQ passive mixer 400 reduces distortion introduced by the source follower NMOS transistors 440, 444, 448 and 452; however the source follower transistors are no longer linear. This limits the linearity of the IQ passive mixer 400.
For a mixer performing an up-conversion frequency translation, a typical specification used is called FRF-3BB (Delta). This is the ratio of the up-converted RF signal to the third order distortion, where the third order distortion is FLO-3.FBB (FLO is the local oscillator frequency and FBB is the frequency of the baseband input signal). For a 2G application, a typical Delta of 55 dB is required. For a 3G voice application, a typical Delta of 45 dB is required.
Thus, to have high Delta, the source follower NMOS transistors 440, 444, 448, and 452 shown in FIG. 4 are required to have large transconductance (gm). The transconductance (gm) for a source follower transistor is directly proportional to the drain current ID of the source follower transistor. Therefore, in order to achieve a high delta value, the current consumption of the source follower transistor must also increase.
The transconductance gm varies with the baseband input signal, due to resulting variations in the drain current. To minimize the effect of the variations, additional resistors 460, 462, 464, and 468 are added in series with the inherent (1/gm) resistance of the source follower NMOS transistors (440, 444, 448, and 452) to improve the linearity of the IQ passive mixer 400.
One trade-off with this design is the value of the resistance of resistors 460, 462, 464, 466 and the Delta value. With a high resistor value, the Delta value increases, however the SNR decreases. Similarly, with a low value resistor the SNR increases, however the Delta value decreases.
It is an aim of the present invention to provide a driver circuit for a mixer circuit with improved linearity.